Global loop integrating modulator

ABSTRACT

A switch-mode modulator operating at a two-level voltage and including an alternating output stage ( 3 ), an optional output filter ( 5 ) and a feedback including a function block ( 14 ) with a transfer function (MFB). The modulator furthermore includes a forward block ( 12 ) provided with means for calculating the difference between the signal ( 14   a ) originating from the function block ( 14 ) and a reference signal (REF) as well as with a transfer function (MFW). The output ( 13 ) of the forward block ( 12 ) is the input of a Schmitt-trigger ( 9 ), which generates switch on signals ( 2 ) for changing the output stage. The output voltage (V OUT ) of the modulator applying either after the optional output filter ( 5 ) or the output stage ( 3 ) is fed back through the function block ( 14 ) so as to generate the signal ( 14   a ) fed back. The transfer function of the function block ( 14 ) and of the forward block ( 12 ) is chosen both in response to the transfer function of the output filter ( 5 ) and in response to the desired total open-loop-transfer function of the modulator.

TECHNICAL FIELD

The invention relates to a switch-mode modulator as described in thepreamble of claim 1.

BACKGROUND OF ART

Power amplifiers are well-known components used for amplifying electricsignals at the signal level, viz. signals at a usually low voltagelevel. Such a low voltage level can for instance be ±2 volt, ±15 volt or0 to 5 volt, i.e. the voltages at which ordinary electronic circuitsoperate. The power amplifier serves to amplify such a low voltage up toa considerably increased power potential. The latter power potential isobtained by intensifying the low voltage to an increased voltage, i.e.by upscaling the low voltage with the result that the signal level isseparated from the power level.

Usually, the power amplifiers are classified in class A, class B, classAB, class C and class D amplifiers. A power amplifier classified inclass A, class B, class AB or Class C is based on the linear, activearea of a transistor. Therefore, the power amplifier can be consideredan ideal, controlled voltage source connected in series to an internalresistance, the load of said power amplifier being connected in seriesto said internal resistance. Elementary circuit calculations show that arather significant portion of the power supplied by the ideal voltagesource is allocated to the internal resistance and thereby not to theload. In theory, it is possible to achieve a maximum efficiency, i.e.the relation between the power received and the power delivered by thepower amplifier, of almost 80% by means of a class B amplifier, but inpractice, the efficiency does not usually exceed 60%. The power loss ismainly allocated in the power amplifier in form of heat, and said heatmust be carried away such as for instance by way of cooling. Usually,the voltage source is not ideal, and it depends on the desired powerlevel, the extent and character of the load as well as on thefrequencies involved.

When an input signal with a known frequency content is compared with theresulting output signal, and when said output signal is subjected to afrequency analysis, differences are inevitably found in the relationbetween the amplitudes of the individual frequencies and the frequenciesfound in the signal. In addition, these relations change in response tothe load on the power amplifier. Therefore, it is possible to indicate afigure for the distortion of the power amplifier versus the frequencyand the power, and this figure represents a quality figure for the poweramplifier which can be used for determining the class of said poweramplifier.

The power amplifiers of the classes A, B, AB and C have been constructedfor many years, and the advantages and limitations thereof arewell-known as well. However, the power amplifiers of the class D differfrom the other power amplifiers by the output transistors thereof beingused as switches or electronic switch elements. Most of the internallosses found in a linear amplifier are caused by the fact that theoperating point of the transistors used is found in the linear area.This linear area shows a rather significant potential drop across thetransistors while a rather significant current passes through said area.On the other hand, when the transistors are always “switched on”, i.e. alow potential drop applies across the transistors while the currentthrough said transistor differs from 0; or when said transistors are“switched off”, i.e. the potential drop across said transistor issignificant while the current through the transistor is very low, almost0; then the resulting power loss in the transistors is always low.Furthermore, the only period involving a significant power loss is theperiod during which the transistor changes from being switched on tobeing switched off, i.e. when both the current and the voltage differfrom zero. Now, when the transistor is switched on and off at a highspeed, viz. at a high frequency, the resulting voltage is a squarevoltage. When the relation between the switched on and off periods ofthe transistors is varied, the continuous average value of the resultingvoltage represents a predetermined value. The latter pre-determinedvalue can be controlled in such a manner that it is possible to ensurethat said value corresponds to the input signal to the power amplifier.Accordingly, it has now been rendered possible to structure a poweramplifier presenting a significantly improved efficiency compared to thehitherto known linear amplifiers.

A power amplifier operating according to the above principle is oftencalled a switch-mode power amplifier or a pulse width-modulated poweramplifier. Such an amplifier implies that a control signal must begenerated for the transistors in order to switch said transistors on andoff in response to the input signal. The control signal is for instancegenerated by a comparison of the input signal with a triangular voltage,and when the triangular voltage presents a value lower than the inputvoltage, a control signal is generated for the transistors reading thatsaid transistors must either be switched on or off. When the triangularvoltage assumes a value higher than the value of the input signal, asecond control signal is generated for the transistors, said secondcontrol signal presenting a value opposite the value of the precedingsignal. The frequency of the triangular voltage is often called thechange-over frequency or the switch-frequency. This switch-frequencydepends inter alia on the intended use of the power amplifier. When itis a question of a class D amplifier used for playing for instancemusic, an advantage is found in using the frequency characteristics ofthe human ear as a starting point when the frequency is to be chosen.The maximum audible frequency for the human ear is usually approximately20 kHz, and accordingly an advantage is found in choosing aswitch-frequency which is considerably higher than said frequency, as anaudible howl, a humming or hissing sound is otherwise found in theresulting sound reproduction. Therefore, the resulting output voltage isa box-shaped or pulse-shaped voltage where the width of the individualpulses varies in response to the input signal.

Optionally, the output voltage (scaled down) and the input voltage maybe input to an integrator, which integrates the voltage differencebetween the two signals. The resulting voltage signal is a triangularwaveform with a changing frequency. This signal is then fed to acomparator with hysteresis. The voltage from the comparator is a squarewaveform, where the pulse width is dependent on the input voltage. Theadvantage is that no additional dedicated circuitry is required forproducing the triangular waveform as described in the previous section.The frequency of the waveform, however, is dependent on the inputvoltage and may become very low, when the input voltage is near or atmaximum.

Usually, the switch-frequency used is considerably higher than thefrequency maximum to the human ear, and accordingly thisswitch-frequency does not usually result in audible nuisances. However,the rapidly alternating current and voltage can easily cause aradio-frequency interference, and therefore it is often necessary toinclude a filter in order to avoid such an interference. Often a secondorder filter is used where an inductance is coupled in series to theoutput of the power amplifier and where a capacity is connected to theend of the inductance adjacent the output and to the frame reference.Therefore, the resulting output voltage complies rather well with theinput signal.

However, the output filter involves complications. In particular, whenthe power amplifier is unloaded, the resonance frequency of the filtercan be excited, and as a result of the poor attenuation of the filter,viz. a high Q-factor, a rather significant voltage rise can be appliedto the output. Such a voltage rise is problematic as it can causedamages to the power amplifier. This problem has often been solved byslightly oversizing the components of the power amplifier and/or byproviding the output of said power amplifier with a load resistor. Thisload resistor is permanently coupled between the positive and negativepoles of the output with the result that it provides an attenuation ofthe resonance of the output filter. This load resistor must necessarilybe of such a size that it is able to efficiently attenuate the resonanceof the output filter without causing a too high additional loss in thepower amplifier. The load resistor is usually a cost-intensivecomponent, and accordingly an advantage is found in avoiding the use ofsuch a load resistor. Usually, the power amplifier is not used in theunloaded state, and therefore the use of a load resistor in form of aloading impedance, such as an RC-element with a high resistance,minimizes the power loss.

As mentioned above it is necessary to use an output filter in order tomake the input signal comply well with the resulting output signal.However, the output filter is not ideal, and the components of thefilter may present non-linear properties depending on temperature,frequency, current etc. Therefore, the output filter causes often per sea distortion of the desired output signal.

It is possible to considerably reduce a number of the draw-backsassociated with a conventional linear amplifier by using a class Damplifier, but such a class D amplifier does per se also involveproblems raising the price of the power amplifier and causingundesirable characteristics, such as noise from the switch-frequency,distortions from the output filter, overvoltages etc.

DE-PS No. 198 38 765 discloses a power amplifier employing a hysteresiscontrol for generating pulse width-demodulated voltages. The differencebetween the input voltage and the output voltage is integrated in thispower amplifier, said difference being stepped down by a factorcorresponding to the ratio of the maximum level of the input voltage tothe maximum level of the output voltage. The difference between thescaled output signal and the input signal corresponds to theinstantaneous amplitude error of the output signal with the result thatthe integration corresponds to the accumulated error on the output. Theoutput signal of the integrator is triangular, and when the poweramplifier is idle running, the slope of said triangle is of the samevalue for both the positive and the negative flanks. When the poweramplifier is to be set, i.e. loaded, these flanks change in such amanner that the positive flank discloses a slope differing from theslope of the negative flank. However, the curve shape remains triangularwith straight flanks. As the power amplifier is increasingly loaded, theswitch-frequency decreases as well. As a result, for instance the inputsignal to the power amplifier is sinusoidal, and then theswitch-frequency is at maximum at the zero-pass for the sine curve andsignificantly lower at the maximum and the minimum value, respectively,of said sine curve. When the power amplifier is loaded to its maximum,i.e. when the maximum value of the output voltage is almost identicalwith the internal DC-voltage of the power amplifier, then theswitch-frequency becomes very low, almost zero. The triangular signalfrom the integrator is transferred to a comparator, typically aSchmitt-trigger, which converts the triangular signal into square pulsesof a varying pulse width. These square pulses are the switching onsignals and the switching off signals, respectively, for the transistorsin the power amplifier. These switching on pulses are transferred to theoutput stage of the power amplifier, viz. to the transistors in theoutput, and therefore these pulses are upscaled by the relation betweensaid pulse voltages and the internal DC-voltage of the power amplifier.The resulting voltage includes square pulses and is typically of ahigher amplitude than the signal voltage. The square voltage is thentransmitted to the output filter of the power amplifier, said outputfilter typically being a second order filter which is often referred toas a reconstruction filter. The voltage applying after the filter is theoutput voltage of the power amplifier. The voltage returned to theintegrator is the voltage applying before the output filter. A modulatorof this type is often referred to as an Astable Integrating Modulator oran AIM. Such a modulator is encumbered with the problem that thedistortions of the output filter have not been taken into account. Inaddition, the operational amplifier used to construct the integrator hasto be of high quality.

WO 02/25357 discloses a controlled oscillation modulator, also called aCOM. The COM ensures that the open-loop-phase characteristics involve aphase shift of exactly 180° at the frequency where theopen-loop-amplification is 0 dB. The latter is rendered possible by thefeedback voltage from the output stage of the power amplifier beingforwarded through function blocks causing a phase shift of 180° and/orthrough function blocks with time delays. The desired phase shift of180° is obtained by including said phase shift in the function blocks,such as in form of a cascade coupling of poles, and/or by choosing asuitable time delay. When the feedback loop is subsequently closed, themodulator oscillates at the frequency where the amplification is 0 dB.When the input signal to the power amplifier is 0, the resulting signalis a substantially pure sine. When the input signal differs from 0, theoscillation is superimposed by the input signal. A comparator issubsequently used for generating the switching pulses of the outputstage. An increasing loading of the amplifier has the effect that thepure sine resulting from the phase shift of 180° is altered into beingsomething between the pure sine and the triangular voltage known fromAIM. The linearity of a modulator depends on variations in theinclination of this signal. As this signal is not a pure triangularcurve unlike AIM, but instead something between a sine curve and atriangular curve, the modulator according to the COM principle isnonlinear, and the modulation per se distorts the output signal.

WO 98/19391 describes a way of improving a power amplifier of the Dclass. The amplifier includes an internal modulator generating thewell-known pulse-width-demodulated output signal. This signal istransmitted to an output filter, and the resulting filtered signal isthe output voltage of the power amplifier. In order to compensate forthe distortions of the filter, additional feedback loops have beenincluded, and the characteristics of these feedback loops can compensatefor the distortions of said output filter. The described system includesseveral cascade-coupled feedback loops for compensating the distortions.The system shows an improved procedure structure with respect to poweramplifiers without such feedbacks, but the system is per se very complexand requires much design work in order to achieve the desired effect. Asystem of this type is often referred to as being Multivariable EnhancedCascade Controlled or MECC.

U.S. Pat. No. 6,249,182 B1 discloses a modulator with an outer feedbackloop after the output filter. The feedback has a lag-lead characteristicwhere the combination of the feedback, the output filter and the forwardblock creates a pole at zero, a double pole at the filter frequency anda zero followed by a pole in the feedback block.

U.S. Pat. No. 5,606,789 discloses a tracking converter comprising twoBuck-converters. A discharge element ensures that the two converters aresynchronized. The feedback is a current feedback, and the converteroperates as a voltage controlled current generator.

U.S. Pat. No. 6,489,841 B2 discloses a switch made power amplifier, inwhich a resistor is placed in series with the capacitor of the outputfilter. This results in an output filter with two poles and a zero whichreduces the suppression of noise. Furthermore, the power amplifier isAC-coupled and thus has poor amplification at low frequencies, Also, thepoles f the output filter are far from the zero in the feedback and as aresult, only one pole is chieved for the output filter.

U.S. Pat. No. 6,552,606 B1 discloses a modulator in which the feedbackis the current measured through the capacitor of the output filter. Thepower amplifier is thus a voltage controlled current generator.

WO 03/090343 A2 discloses a power amplifier with a lead-lag feedback.The lead-lag in the feedback results in a second order response for thehigh frequencies.

DESCRIPTION OF THE INVENTION

Accordingly, the object of the invention is to provide a power amplifierof a simple structure, where the modulator per does not causesignificant distortions, and where the distortions of the output filterhave been taken into account and where the closed loop characteristicsof the power amplifier are approximately a first order system. Thisobject is obtained by means of the features described in thecharacterising clause of claim 1. As the output signal is the signalbeing fed back and not the output voltage of the output stage, theeffect of the output filter on the output voltage has been compensatedfor. In addition, the resulting modulator is per se linear, i.e. theoutput voltage of the integrator block includes straight positive andnegative flanks. In other words, the modulator does not per se distortthe output signal. Furthermore, as the output voltage is fed backthrough the function blocks including the transfer function, and as theintegration block can include transfer functions, it is renderedpossible to completely or partially compensate for the zeros of theoutput filter. As described above, an output filter, such as forinstance a second order filter, can be assumed with the result thatrather significant voltage rises can apply to the output of theswitch-mode power amplifier, which in turn can cause damages to eithersaid switch-mode power amplifier or to the equipment coupled to saidswitch-mode power amplifier. As the output voltage after the outputfilter is fed back through the function blocks, it is possible toefficiently compensate for voltage rises on the output. Subsequently, itis not necessary to include an additional load impedance, such as forinstance an RC-element or similar element, on the output of theswitch-mode power amplifier for attenuating or removing theseovervoltages. Therefore it is not necessary to include additionalfeedbacks in order to compensate for the effect of the output filter onthe output voltage. Measurements on a switch-mode power amplifieraccording to the invention have demonstrated that said power amplifierpossesses good properties with respect to distortion of the outputsignal both in relation to the output supplied and in relation to theoutput frequency.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention is explained in greater detail below with reference to theaccompanying drawings, in which

FIG. 1 shows the output stage of a switch-mode power amplifier as wellas said output stage with a reconstruction filter included,

FIG. 2 shows a way of generating PWM-voltages,

FIG. 3 is a diagram of the operation of a switch-mode power amplifier,

FIG. 4 shows a switch-mode power amplifier operating according to theAIM principle,

FIG. 5 shows the resulting modulation voltage of the switch-mode poweramplifier shown in FIG. 4,

FIGS. 6, 7 and 8 show various embodiments of a modulator according tothe invention as well as additional feedback loops,

FIGS. 9 to 20 show various embodiments of a modulator according to theinvention as well as the resulting open-loop-function, and where FIGS. 9to 14 are without output filters and FIGS. 15 to 20 are with outputfilters,

FIG. 21 shows an embodiment of the modulator according to the invention,

FIG. 22 shows the definition for the open-loop-function,

FIG. 23 shows the interaction between the open-loop-function of themodulator and the open-loop-function of the feedback,

FIG. 24 shows how two modulators according to the invention can be usedfor generating three-level voltages,

FIG. 25 shows a way of structuring a Schmitt-trigger, and

FIG. 26 shows how two Schmitt-triggers shown in FIG. 25 can besynchronized.

BEST MODE FOR CARRYING OUT THE INVENTION

In the following description the transistors are illustrated by means ofa simple switch symbol, both because the various types of relevanttransistors are usually illustrated by means of different symbols, andbecause the object is to provide a transistor functioning in a waysimilar to an ideal switch in connections with a switch-mode poweramplifier. The structuring of a switch-mode power amplifier according tothe invention requires, of course, the use of driving circuits for theindividual transistors. For the sake of clarity, these driving circuitshave not been illustrated, and furthermore they are well-known to aperson skilled in the art. It is necessary to post-process a controlsignal generated by for instance a modulator, and such a post-processingcan for instance be carried out by means of so-called dead timegenerators as well as other circuits necessary for driving thetransistors in the output stage of the power amplifier. These circuitsare also well-known to a person skilled in the art, and for the sake ofclarity they are not illustrated.

FIG. 1A shows the most simple form of a switch-mode power amplifier. Asuitable double DC voltage UDC supplied by the power supply of aswitch-mode power amplifier is fed to the illustrated output stage. Thecontrol signals for the electronic switch elements S₁, S₂ determinewhether either the electronic switch element S₁ is closed or theelectronic switch element S₂ is open with the result that the load Z₁ iscoupled between the positive and negative pole of the supply voltageU_(DC); or in the opposite case where the electronic switch element S₁is open and the electronic switch element S₂ is closed, then the load Z₁is short-circuited, i.e. no voltage is applied thereon. The switchelements S₁, S₂ are usually controlled by means of a single controlsignal, where said control signal has the effect that the electronicswitch element S₁ is closed while the electronic switch element S₂ isopened and vice versa. The resulting voltage across the load Z₁ istherefore a square, pulse-shaped voltage. As mentioned above, a poweramplifier does not usually cause audible nuisances due to the switchfrequency, as said switch frequency is usually significantly increasedabove the audible area. However, the filter is necessary for limitingthe radio-frequency interference. Therefore, it is ordinarily desirableto filter out the voltage originating from the output stage of theswitch-mode power amplifier as shown in FIG. 1B. The circuit of FIG. 1Bcorresponds to the circuit of FIG. 1A apart from the fact that thecapacity C1 and the inductance L1 form a filter for the output voltage.A filter arranged in this manner is often referred to as a restructuringfilter. Usually, a restructuring filter includes passive components,such as capacities, resistances and inductors, and such a filtersmoothens the square pulse voltage from the output stage. In addition,such a filter may cause a distortion of the resulting output voltage ofthe switch-mode power amplifier.

FIG. 2 shows one way of generating the output voltages of a switch-modepower amplifier by means of a triangular voltage SW with identical,straight flanks, i.e. both positive and negative. The frequency of thetriangular voltage SW is usually significantly higher than the maximumfrequency to be amplified, typically 10 to 20 times higher or more, suchas in the situation where the power amplifier is used for playing musicwhere the frequency is usually 40 times higher. The latter triangularvoltage is compared with a reference voltage REF, i.e. the input voltagefor the switch-mode power amplifier. As long as the reference voltageexceeds the triangular voltage SW, the resulting output signal from themodulator is “high”, i.e. a signal is generated which switches on theelectronic switch element S₁ and switches off the electronic switchelement S₂ shown in FIG. 1A. At the time T₁, the triangular voltage SWcrosses the reference voltage with the result that the modulatorgenerates a “low” signal, viz. a signal switching on the electronicswitch element S₂ and switches off the electronic switch element S₁shown in FIG. 1A. At the time T₂, the reference voltage REF crosses thetriangular voltage SW again with the result that the modulator generatesa high signal again. Such a procedure continues at all the times wherethe reference voltage REF crosses the triangular voltage SW, i.e. at thetimes T₃, T₄, T₅, T₆, T₇, T₈ and T₉. The reference signal REF shown inFIG. 2 is slowly increasing. When the length of the voltage pulsegenerated between the time T₂ and T₃, viz. the pulse P₁, is comparedwith the length of the pulse generated between the times T₈ and T₉, viz.the pulse P₂, it appears that the pulse length of P₁ is significantlyshorter than the pulse length of P₂, which agrees with the referencesignal REF between the times T₂ and T₃ being lower than between thetimes T₈ and T₉. The triangular signal SW is often referred to as thesignal at a specific frequency, i.e. the switch frequency or the carrierfrequency. The illustrated way of generating the switching on pulses forthe electronic switch elements S₁, S₂ is only one way out of manypossible ways. The triangular voltage can for instance be a sawtoothvoltage. It is known to a person skilled in the art that the use of asawtooth voltage instead of a triangular voltage ensures the sameresult, but it is necessary to frequently switch on and off theelectronic switch elements. The above pulse-formed voltage is atwo-level pulse-width-modulated voltage. Three-levelpulse-width-modulated voltages are also known where the voltagealternates between a zero-voltage and the positive or the negativesupply voltage, respectively. The three-level pulse-width-modulatedvoltages require the use of a means differing from the means shown inFIGS. 1A and 1B, but such a means is also well-known to a person skilledin the art.

FIG. 3 shows a flow chart for a switch-mode power amplifier. Thereference signal REF is compared with the triangular voltage SW by meansof a comparator 1. The function of the comparator 1 corresponds to theway the switching on signals are generated as shown in FIG. 2. Theswitching of signals 2 are transferred to the output stage 3 of theswitch-mode power amplifier, and the resulting square voltage 4 presentsusually a higher amplitude than the switching on signals 2. Theseswitching on signals 2 are transmitted to an output filter 5 with theresult that the desired output voltage V_(OUT) is subsequentlygenerated. Therefore, the function of the illustrated flow chart issubstantially the same as the function illustrated in FIGS. 1 and 2. Theillustrated reference signal REF is shown with a frequency almostcorresponding to the triangular voltage SW, but as mentioned above thefrequency of the triangular voltage SW is usually significantly higherthan the maximum frequency of the reference signal to be amplified, i.e.usually at least 10 to 20 times higher. The frequency of the triangularvoltage is usually kept substantially constant and is generated on thebasis of the oscillations of for instance a crystal or by means of an astable multivibrator. An a stable multivibrator is most preferredbecause usually a very accurate switch frequency is not necessary. Thefrequency of the triangular voltage SW is therefore well-known andalways the same.

FIG. 4 shows a switch-mode power amplifier operating according to theAstable Integrating Modulator principle described in the DE-PS No. DE198 38 765. This principle is also referred to as AIM. As illustrated inFIG. 4, a switch-mode power amplifier operating according to theAIM-principle generates switching on pulses 2 which are transferred tothe output stage 3 of a switch-mode power amplifier. These pulses 2generate square voltage pulses 4 which are transmitted to an outputfilter 5 reconstructing the desired output signal V_(OUT). The lattercorresponds to the general principle illustrated in FIG. 3. However, theway of generating the switching of signals 2 differs from the way shownin FIG. 3. The generated square voltage 4 is transferred to a scalingblock which scales the amplitude of the square voltage 4 by a factor K.This factor K can for instance be the relationship between the maximumamplitude of the reference voltage REF and the maximum amplitude of thesquare voltage 4. The difference between the reference signal REF andthe downscaled square voltage 4, viz. the error signal, is integrated inan integration block 6. The resulting triangular voltage 8 presentsnumeric, straight flanks when the switch frequency is high compared tothe input frequency, but the slope of the positive flank differs fromthe slope of the negative flank. The triangular voltage 8 is transferredto a Schmitt-trigger, i.e. a comparator with a built-in hysteresis. Theoutput voltage resulting from the Schmitt-trigger is in form of thesquare switching on signals 2 for the output stage 3 of the switch-modepower amplifier. As a feedback exists from the square voltage 4, it istherefore possible to ensure that the resulting output voltage V_(OUT)presents an improved agreement with the reference signal, i.e. the formof the output signal is substantially similar to the reference signal.In addition, the distortion of the output signal has been reduced.

FIG. 5 shows both a sinusoidal reference signal REF and a triangularvoltage signal corresponding to the triangular voltage signal 8 of FIG.4. As illustrated, the frequency of the triangular voltage 8 is notconstant, but depends on the loading of the switch-mode power amplifier,i.e. whether the resulting output voltage is high or approximately 0. Itappears from the 0-crossing 11 of the sinusoidal reference voltage REFthat the frequency is high whereas said frequency is significantly lowerat the maximum and minimum value 10, respectively, of the sine curve.The latter is inter alia a consequence of the fact that a switch-modepower amplifier of this type does not employ for instance a crystal formaintaining a specific frequency as described in connection with FIG. 3.Under specific circumstances and/or as a consequence of an unfortunatedesign, the resulting drop in frequency can be so significant that itaffects the audible quality of the output signal.

FIG. 6 shows a switch-mode power amplifier according to the invention.Like the previous examples, this power amplifier employs a switch onsignal 2 for the output stage 3 of the power amplifier, said switch onsignal generating a square voltage 4 being transmitted to an optionaloutput filter 5 which reconstructs the desired output signal V_(OUT).Unlike the AIM described in connection with FIG. 4, the output signalV_(OUT) and not the square voltage 4 is fed back after the output stage3 of the switch-mode power amplifier. The square voltage 4 or thevoltage after the optional filter 5 is fed to a block 14 including thetransfer function MFB for the feedback of the modulator. After the block14, the signal is transmitted to a block 12 including the transferfunction MFW for the forward of the modulator as well as the extractionof the difference between the reference signal and the signal fed back.However, the output signal 13 of the block 12 is instead a triangularsignal with numeric, equally steep flanks, and the positive and thenegative flanks present differing slopes. This signal 13 is transferredto a block 9 including a Schmitt-trigger, and the output signal 2 of theblock 9 is the switch on signal 2 for the output stage of theswitch-mode power amplifier. As a result of structuring the switch-modepower amplifier according to this principle it is allowed to take intoaccount the distortions of the output filter 5 and the output filter perse. In addition, it is possible to ensure that the modulator does notper se cause a substantial distortion of the output signal V_(OUT) dueto the straight flanks of the signal 13.

FIG. 6 additionally shows a plurality of optional components indicatedby means of a dotted line. These optional components can for instance beadditional output filters 15, one or more additional feedbacks 16 withtheir respective transfer functions CFB_(1-N) and one or more forwardblocks 17 with their respective transfer functions CFW_(1-N). Asindicated by means of the dotted line 18, the output filters 5, 15 canfor instance be connected and disconnected to the system and thereby“short-circuit” said system. FIGS. 7 and 8 show variations of theembodiment shown in FIG. 6, where the embodiment of FIG. 7 only includesone additional block 16 in the feedback, and where the forward blocks 17of the embodiment of FIG. 8 use the same feedback 14 as the modulatoraccording to the invention. The modulator according to the inventionwill in the following be referred to as a GLIM-type modulator (Globalloop integrating modulator).

Therefore, the amplifier or power supply according to the inventionachieves the advantages through the structure of the modulator.Therefore, the open-loop-characteristics of a modulator according to theGLIM-principle are an approximation to an integration, i.e. a pole inzero.

Modulators according to the GLIM-principle can be divided up into twomain families:

-   -   1. one type of modulators operating around the output stage        only, and    -   2. one type of modulators operating around the output stage and        the output filter.

A characteristic frequency can be used for both types of modulators. Asfar as type 1 and type 2 modulators are concerned, the frequencydiffering from zero forms part of a pole or zero in one or more of theblocks of the modulators. As far as type 2 modulators are concerned, thecharacteristic frequency is often chosen so that it coincides with thefilter frequency of the output filter used. As far as type 2 modulatorsare concerned, the characteristic frequency is then the same as thepower bandwidth.

Each of the two main families can be subdivided into two sub-families:

-   -   1. modulators including a single loop, i.e. a pure modulator,        and    -   2. modulators including several loops (a modulator and a        feedback loop or loops).

In addition, the realization of GLIM including a modulator loop with orwithout output filter and combined with one or more feedback loops withor without further output filters can be divided up into twosub-families:

-   -   1. modulator loops with pure first order low-pass        characteristics at frequencies higher than the characteristic        frequency, and    -   2. modulator loops with partial first order low-pass        characteristics at frequencies higher than the characteristic        frequency, such as layer-lead-characteristics with pole        frequency in the characteristic frequency.

When the modulator loop does not present pure first ordercharacteristics, an associated feedback circuit must ideally be formedso that the modulator and the feedback together present the desiredfirst order low-pass characteristics. However, in practice, GLIM can beconstructed so as to include a deviation therefrom as GLIM per seincludes the modulator and only indicates guidelines for possibleadditional feedback circuits. Type 1 U/Output filter Type 2 M/Outputfilter Type 1.1 Type 1.2 Type 2.1 Type 2.2 Pure modulator (AIM) Combinedmodulator and feedback Pure modulator Combined modulator and feedbackType 1.2.1 Type 1.2.2 Type 2.2.1 Type 2.2.2 Modulator with firstModulator deviating Modulator with first Modulator deviating orderlow-pass from first order low- order low-pass from first order low-characteristics pass characteristics characteristics passcharacteristicsBasic Properties of the Types

Type 1) Modulator operating around output stage

-   -   Ideally, the feedback and the forward block must together form a        first order low-pass-characteristic at frequencies higher than        the desired bandwidth.

Type 2) Modulator operating around the output stage and the outputfilter.

-   -   Ideally, the output filter, the feedback and the forward block        must together form a first order low-pass-characteristic at        frequencies higher than the desired band-width. In other words,        when a second order output filter is used, such as for instance        an LC filter, the feedback and the forward block must ideally        together form a zero at frequencies higher than the filter        frequency.        Basic Requirements Presented to the Sub-Families

Type 1)

-   -   Type 1.1        -   Ideally, the feedback and the forward block must together            present an open-loop-characteristics as a pure integration,            i.e. a pole in zero.        -   The modulator is an independent modulator not requiring            further adjustment of the linear function.    -   Type 1.2        -   Ideally, the total open-loop-characteristics for the            modulator and the feedback loop must be a pure integration,            i.e. a pole in zero.        -   Ideally, the open-loop-characteristics of the feedback loop            must be the difference between the desired pure integration            and the open-loop-characteristics of the modulator.            -   Type 1.2.1                -   Ideally, the modulator presents a pure first order                    open-loop-low-pass-characteristic at frequencies                    above the filter frequency, i.e. when the                    open-loop-characteristics of the modulator is a                    constant amplification by means of a pole, then the                    feedback loop(s) must ideally present an                    open-loop-characteristic as a pole in zero as well                    as a zero in the pole frequency of the modulator.            -   Type 1.2.2                -   The modulator deviates from presenting a pure first                    order open-loop-low-pass characteristic at                    frequencies above the filter frequency. The                    modulator can for instance present a lag-lead                    characteristic, which then ideally requires an                    open-loop-characteristic for the feedback loop(s) in                    form of a pole in zero, a zero in the pole frequency                    of the modulator as well as a pole in the zero                    frequency of the modulator.

Type 2)

-   -   Type 2.1        -   Ideally, the output filter, the feedback and the forward            block must together present an open-loop-characteristic as a            pure integration, i.e. a pole in zero.        -   The modulator is an independent modulator not requiring            further adjustment of the linear function.    -   Type 2.2        -   Ideally, the total open-loop-characteristics for the output            filter, the modulator and the feedback loop must be a pure            integration, i.e. a pole in zero.        -   Ideally, the open-loop-characteristics of the feedback loop            must be the difference between the desired pure integration            and the open-loop-characteristics of the modulator.        -   Type 2.2.1            -   Ideally, the modulator presents a pure first order                open-loop-low-pass-characteristic at frequencies above                the filter frequency, i.e. when the                open-loop-characteristics of the modulator is a constant                amplification by means of a pole, then the feedback                loop(s) must ideally present an open-loop-characteristic                as a pole in zero as well as a zero in the pole                frequency of the modulator.        -   Type 2.2.2            -   The modulator deviates from presenting a pure first                order open-loop-low-pass characteristic at frequencies                above the filter frequency. The modulator can for                instance present a layer-lead characteristic, which then                ideally requires an open-loop-characteristic for the                feedback loop(s) in form of a pole in zero, a zero in                the pole frequency of the modulator as well as a pole in                the zero frequency of the modulator.

The open-loop-transfer function of a modulator according to theGLIM-principle depends inter alia on the modulator-feedback MFB chosenfor the block 14, and on the transfer function MFW chosen for theforward block 12.

FIG. 9 shows how the transfer functions MFB, MFW are chosen in generalin the feed-back block 20 and the forward block 19, which results indiffering open-loop-functions 21.

FIGS. 10, 11, 12, 13 and 14 show five possible examples of resultingopen-loop-functions for GLIM-modulators of the first main family.However, the examples shown are not to be considered limited to theseexamples as several other possibilities exist.

FIG. 10 shows a specific embodiment corresponding to AIM.

FIG. 15 shows a GLIM-modulator of the second main family, i.e. amodulator operating both as output stage and as output filter 22.

FIGS. 16, 17, 18, 19 and 20 show various possible transfer functions forthe blocks 19 and 20 and the resulting open-loop-function 21. It shouldbe noted, that the illustrated examples are nothing but a few examplesout of many possible examples.

FIG. 21 shows a particularly simple way of structuring a modulatoraccording to the GLIM-principle. In this embodiment, the output voltageV_(OUT) is fed back to a first impedance, such as a parallel connectionof a resistance K×R and a capacitor C. The reference signal REF isapplied a summation point (SUM) through a second impedance, such as aresistance R, and after the summation point (SUM) the voltage V₁ istransferred to the Schmitt-trigger 9. As illustrated in the example,such a procedure provides two bends in the transfer function for theopen-loop-function, i.e. at a frequency determined by K×R and afrequency determined by R. As shown, the components of the feedbackcircuit ensures both the desired zero in the filter frequency, but alsoa pole at the frequency K× of the filter frequency, where K is theLF-amplification of the modulator. However, it is easy to compensate forsuch a deviation by allowing the output of the feedback circuit to havea zero at this frequency. Such a technique is described below. Anadvantage of the two types of modulators 1.2.x and 2.2.x is that themodulator per se can be formed by means of passive components, which inturn is both simple and ensures the advantage that no active weak signalcomponents must include the sawtooth carrier wave signal. As a result,the HF-performance requirement presented to the active weak signalcomponents has been considerably reduced. As a result, it is possible touse inexpensive types of components for this purpose, and theperformance of the weak signal components in the frequency area inquestion for the power supply/the amplifier is not significantly reducedas a consequence of a non-linear function caused by an HF-content in thesignal to be processed by said components.

By optionally using several feedback loops, the total suppression oferrors originating from the output stage as well as from the outputfilter can be increased significantly, viz. by a function as pole inzero and a zero in the filter frequency resulting in an amplification ofzero-dB at high frequencies.

As mentioned above, a modulator according to the GLIM-principle can beconstructed so that the open-loop-characteristics of the modulator loopper se deviates from being an approximated pure integration. Ideally,the requirement presented to the associated feedback system is that theopen-loop-characteristics thereof are the difference between the pureintegration and the open-loop-characteristics of the modulator.

FIG. 22 shows the definition used for the interrupting locations of theopen-loop-characteristics for the modulator and the feedback circuit.Accordingly, the open-loop-characteristics of the modulator isL_(modulator)=W_(M2)/W_(M1) and correspondingly for the feedback circuitby means of one or more feedback loops L_(control)=W_(C2)/W_(C1). Pleasenote the definition in the figure of W_(M1), W_(M2), W_(C1), W_(C2),defining where the loop is opened corresponding to FIG. 7.

FIG. 23 shows three examples of open-loop-functions for the modulatorand the associated ideal feedback loops. The open-loop-function of themodulator is indicated by means of a solid line, and the transferfunction of the feedback is indicated by means of a dotted line. Theexample A shows a modulator of type 1.1 or type 1.2. It appears that therequirement to the feedback system is in fact a constant amplificationwhich is due to the fact that the modulator can operate alone becauseinherently it possesses the desired transfer function. Example B and Care examples of open-loop-functions where the modulator is of the typeX.1.2 and the type X.2.2, respectively. It appears from Example B thatthe open-loop of the modulator presents a bend at a predeterminedfrequency, and accordingly the feedback loop has also a bend in theopposite direction at the same frequency. In example C the modulatorpresents two bends at different frequencies, and therefore theopen-loop-transfer function of the feedback must also have twocorresponding bends in the opposite direction.

Above a modulator according to the GLIM-principle has been described fora power amplifier/supply voltage employing two-level voltages, i.e.which either applies its total supply voltage or a zero voltage.However, a power supply or a power amplifier can also employ three-levelvoltages, where either a zero voltage or the positive or negative supplyvoltage, respectively, is applied onto the load. The latter can forinstance be carried out by means of an H-bridge well-known to a personskilled in the art. For this purpose it is necessary to generate twocontrol signals instead of one control signal unlike the abovemodulator, and these two control signals are generated by means of amodulator, cf. FIG. 24, which includes two modulators M1, M2corresponding to the above GLIM-modulator. The same reference signal isapplied onto both modulators M1, M2, but the signal for one modulator M1is inverted by means of an inverter 24. The two modulators M1, M2 areusually identical. In principle, the two modulators M1, M2 can operatecompletely independently of one another as the power supply load of thepower amplifier usually synchronizes the two modulators to apredetermined extent. In practice, it is often necessary to implement asynchronizing circuit between the two modulators, cf the synchronizingblock 23 in FIG. 24. As illustrated, the synchronizing block 23 iscoupled between the Schmitt-triggers of the two modulators M1, M2, butthey can just as well be coupled between other blocks of the modulators,such as for instance the output stage.

The Schmitt-trigger used in the above modulators is a componentwell-known to a person skilled in the art. The output of such aSchmitt-trigger alternates in response to the difference between thevoltages applied onto its input. When one input exceeds the other input,the output is for instance the positive supply voltage; whereas when thesecond input exceeds the first input, then the output is zero or thenegative supply voltage.

In a modulator according to the invention, the Schmitt-trigger canadvantageously be constructed as shown in FIG. 25. In such aSchmitt-trigger, the output voltage V_(OUT) of the output stage isconnected to a first resistance R₁ by means of a second resistance R₂and further to the frame potential. The transition between the tworesistances R₁, R₂ is connected to a summation point, where the secondinput for said summation point is the input voltage of theSchmitt-trigger, and where the output from said summation point is theinput for the output stage. The hysteresis-window V_(SCH) is determinedas V_(SUPPLY)×R₂/(R₁+R₂), where V_(SUPPLY) corresponds to the supplyvoltage for the output stage.

As mentioned above, the two modulators M1, M2 of FIG. 24 aresynchronized, and here it is especially advantageous that the twoSchmitt-triggers are the components to be synchronized. However, othercomponents may be synchronized too. The synchronizing can be carried outin a simple way as shown in FIG. 26, where a resistance R₃ is coupled inthe transition between the two resistances R₁, R₂ of the Schmitt-triggerof the first modulator M1 and in the transition between the tworesistances R₁, R₂ of the Schmitt-trigger of the second modulator M2.The synchronizing is carried out by a minor portion of thehysteresis-window of one modulator is added to the hysteresis-window ofthe second modulator.

The invention has been described with reference to a preferredembodiment. However, the scope of the invention is not limited to theillustrated embodiment, and alterations and modifications can be carriedout without deviating from said scope of the invention. The currentmeasured through the output filter can for instance be used as feed-backsignal instead of the output voltage. As a result, the power amplifieris operated by means of current instead of by means of voltage, wherebythe advantage is obtained that it is not necessary to implement anovercurrent protection in the switch-mode power amplifier. The preferredembodiment employs an output filter in form of a second order filter,but other types of passive filters can be used as well.

1. Switch-mode modulator operating at a two-level output voltage andincluding an alternating output stage (3), an optional output filter (5)and a feedback including a function block (14) with a transfer function(MFB), said modulator further including a forward block (12) providedwith means for calculating the difference between the signal (14 a)originating from said function block (14) and a reference signal (REF)as well as with a transfer function (MFW), where the output (13) of theforward block (12) is the input of a Schmitt-trigger (9), saidSchmitt-trigger generating switch on signals (2) for switching theoutput stage, characterised by feeding back the output voltage (V_(O)UT)of the modulator either after the optional output filter (5) or afterthe output stage (3) through the function block (14) so as to generatethe signal (14 a) fed back, and where the transfer function of thefunction block (14) and of the forward block (12) depends on thetransfer function of the output filter (5) as well as on desired totalopen-loop-transfer function of the modulator when (MFB, MFW) is chosen.2. Modulator according to claim 1, characterised in that one or more ofthe zeros or poles of the function block (14) and of the forward block(12) are chosen to be coinciding or approximately coinciding with thezero points of the output filter (5).
 3. Modulator according to claim 1,characterised by further including one or more additional output filters(I 5), one or more additional feedbacks with transfer functions(CFB_(N)) coupled either after the output stage (3), after the outputfilter (5) or after one or more of the additional output filters (15),as well as one or more forward 25 blocks (17), which include both meansfor calculating the difference between one of the fed back signals fromthe additional function blocks (16) and a reference signal, as well as atransfer function (SFW_(N)).
 4. Modulator according to claim 3,characterised by the transfer function (MFB) 30 of the function block(14), the transfer function (MFW) of the forward block (12), thetransfer functions (CFB_(N)) the additional output filters (15), thetransfer functions (CFW_(N)) of the additional function blocks (16) andthe additional forward blocks (17) together generating the desired totaltransfer function.
 5. Modulator according to claim 4, characterised byincluding only one additional function block (16) used as feedback forone or more additional forward blocks (17).
 6. Modulator according toclaim 4, characterised by using the output signal (14 a) of the functionblock (14) for feeding back to one or more additional forward 10 blocks(17).
 7. Modulator according to claim 1, characterised by aiming atmaking the desired total transfer function being similar to a 1st orderlow-pass-characteristics.
 8. Modulator according to claim 1,characterised by structuring the function block (14) of the feedback andthe forward block (12) in such a manner that the output voltage iscoupled either after the output stage (3) or after the output filter (5)to a summation point (SUM) through a first impedance, especially a 20parallel coupling of a first resistance (K×R) and a capacitor (C), andfurther in such a manner that the reference (REF) is coupled to thesummation point (SUM) through a second impedance, especially a secondresistance (R), that the output voltage (V_(t)) of the summation point(SUM) is coupled to the input of the Schmitt-trigger (9).
 9. Switch-modemodulator operating at three-level output voltages and including twoswitch-mode modulators as claimed in claim 1, characterised in that themodulator includes a first branch with a first switch-mode modulatoroperating at two-level-output voltages (M1) and a second branch with asecond switch-mode modulator operating at two-level-output voltages(M2), where the input signal (V_(JN)) for 30 the first modulator (M1)and the input for the second modulator (M2) are mutually inverted, andthat the voltage between the output (V_(OU)T+) of the first modulator(M1) and the output (V_(QUT)−) of the second modulator (M2) is theresulting output voltage (V_(O)UT).
 10. Modulator according to claim 9,characterised in that a synchronizing unit (23) is coupled between thefirst modulator (M1) and the second modulator (M2) and is adapted tosynchronize the two modulators (M1, M2) to one another.
 11. Modulatoraccording to claim 10, characterised in that the two Schmitt-triggers ofthe two modulators (M1, M2) are synchronized relative to one another.12. Modulator according to claim 1 characterised in, that theSchmitt-trigger (9) is constructed by the voltage after the output stage(3) or the voltage after the output filter (5) being fed back to a firstsummation point (SUM1) through a first resistance (R)]) and furtherthrough a second resistance (R₂) to frame (25), by the input voltage(V_(IN)) being deducted from the voltage from the first summation point(SUM1), and by the output voltage from the second summation point (SUM1)being the input of the output stage (3).
 13. Modulator according toclaim 12, characterised in that the Schmitt-triggers of the twomodulators (M1, M2) are synchronized by a resistance (R₃) being coupledbetween the first, summation point (SUM1) of the first modulator (M1)and the first, summation point (SUM1 ¹) of the second modulator (M2).